The present invention relates to a frequency error estimating apparatus for estimating a frequency error or a frequency differential between a local oscillation frequency and a carrier frequency of a received signal in a receiver used for a satellite communication, a mobile satellite communication and a mobile communication. The invention particularly relates to a frequency error estimating apparatus capable of securing high precision in estimating a frequency error without losing the level of following the time variation in the Doppler frequency.
A conventional frequency error estimating apparatus will be explained below. Some of the receivers used for mobile communications employ a synchronous detection system that is capable of obtaining satisfactory detection characteristics even in a low C/N channel. According to this synchronous detection system, a carrier that is synchronous with a carrier frequency of a received signal is reproduced, and a detection output is obtained based on this carrier. However, the receiver that employs the synchronous detection system has variations in the oscillation frequency due to variations in the precision of the oscillator, variations in temperature, etc. As a result, there arises a difference in the frequency between the transmitter and the receiver. In other words, a frequency error occurs between the transmitter and the receiver. When such a frequency error exists, the phase at a signal point on an IQ plane (a complex plane expressed by a real axis and an imaginary axis) rotates.
Therefore, in order to minimize this frequency error and to improve the synchronous characteristics in the receiver, it becomes necessary to provide a frequency error estimating apparatus that measures a phase rotation volume from a received signal and estimates a frequency error from a result of this measurement.
FIG. 8 shows a structure of a conventional frequency error estimating apparatus. This frequency error estimating apparatus is disclosed in xe2x80x9cDoppler-Corrected Differential Detection of MPSKxe2x80x9d, IEEE Trans. Commun., Vol. COM-37, 2, pp. 99-109, Feb., 1989. In FIG. 8, legend 1 denotes a received signal, legend 21 denotes an M-multiplier for removing a modulation component of the received signal 1, and legend 22 denotes a D-symbol differential detector for performing differential detection over a period of D symbols based on the output of the M-multiplier 21. Legend 101 denotes an averaging filter for averaging the output of the D-symbol differential detector 22, and thereby suppressing a noise component. Legend 24 denotes a coordinate converter for calculating a phase component from the output of the averaging filter 101, and legend 25 denotes a divider for calculating a frequency error from a phase component that has been output from the coordinate converter 24. Legend 2 denotes an estimated frequency error value that is output from the divider 25.
FIG. 9 shows an example of an internal structure of the averaging filter 101. This shows an IIR (a primary infinite impulse response) filter. In FIG. 9, legends 111 and 112 denote multipliers for multiplying an input signal by a specific coefficient respectively. Legend 32 denotes an adder for adding two inputs, and 33 denotes a delay unit for delaying a signal by one symbol.
The conventional frequency error estimating apparatus having the above-described structure is a D-symbol differential detection type frequency error estimating apparatus that estimates a phase change volume due to a frequency error, by performing differential detection over a period of D symbols based on a received signal.
The operation principle of the conventional frequency error estimating apparatus will be explained with reference to FIG. 8 and FIG. 9. When a modulation system used is the M-phase PSK (phase shift keying) system, the received signal 1 (r(nT)) is expressed by the following equation (1).
r(nT)=A(nT)exp[j{xcex8(nT)+xcex94xcfx89nT}]xe2x80x83xe2x80x83(1) 
In the equation (1), the received signal 1 (r(nT)) is a complex base band signal sampled in a symbol period T. A(nT) expresses an amplitude component, and xcex94xcfx89 expresses an angular frequency error. xcex8(nT) expresses a modulation component, and this takes M values of, for example, 2xcfx80k/M (k=0, 1, . . . , and Mxe2x88x921). To simplify the explanation, it is assumed that there is no noise component.
The M-multiplier 21 multiplies the received signal 1 by a modulation multiple number M for removing the modulation component of the received signal 1. A signal after the multiplication (r1(nT)) is expressed by the following equation (2).
r1(nT)=A(nT)exp[jM{xcex8(nT)+xcex94xcfx89nT}]xe2x80x83xe2x80x83(2) 
In the equation (2), Mxcex8(nT) is a multiple of 2xcfx80, and therefore, this can be disregarded. The equation (2) can be substituted by the following equation (3).
r1(nT)=A(nT)exp(jMxcex94xcfx89nT)xe2x80x83xe2x80x83(3) 
The D-symbol differential detector 22 performs differential detection over a period of D symbols based on the output (r1(n)) from the M-multiplier 21. A signal after the differential detection (d1(nT)) is expressed by the following equation (4).
d1(nT)=r1(nt)r1*(nTxe2x88x92DT)=A(nT)A(nTxe2x88x92DT)exp(jMDxcex94xcfx89T)xe2x80x83xe2x80x83(4) 
In the equation (4), r1* (nTxe2x88x92DT) is a conjugate complex number of r1(nTxe2x88x92DT).
The averaging filter 101 averages the output (d1(nT) of the D-symbol differential detector 22, and thereby suppresses the noise component. For example, when the primary IIR filter shown in FIG. 9 is used as the averaging filter, an output (d2(nT)) of the averaging filter 101 is expressed by the following equation (5).
d2(nT)=xcex1d1(nT)+(1xe2x88x92xcex1)d2(nTxe2x88x92T)xe2x80x83xe2x80x83(5) 
In the equation (5), the first term is a result of the multiplier 111 multiplying the input signal (d1(nT)) by the coefficient xcex1, and the second term is a result of the multiplier 112 multiplying the one symbol-delayed output (d2(nTxe2x88x92T)) of the averaging filter 101 by the coefficient 1xe2x88x92xcex1.
When it is assumed that the sampling timing is a Nyquist point, that is, when the amplitude component is assumed as 1, the output (d2(nT)) of the averaging filter 101 is expressed by the following equation (6).
d2(nT)=exp(jMDxcex94xcfx89T)xe2x80x83xe2x80x83(6) 
The coordinate converter 24 converts the output (d2(nT)) of the averaging filter 101 from a Cartesian coordinate into a polar coordinate, and calculates the phase component (MDxcex94xcfx89T). Last, the divider 25 divides the phase component (MDxcex94xcfx89T) that is the output of the coordinate converter 24 by MD, thereby to calculate the angular frequency error (xcex94xcfx89T) over one symbol, and outputs a calculated result.
According to the above-described conventional frequency error estimating apparatus, however, in order to estimate a frequency error in high precision, it is necessary to set the coefficient xcex1 of the multiplier in the averaging filter to a value as small as possible for increasing the averaging effect. On the other hand, when the received signal receives a large Doppler shift and the Doppler frequency further varies with time like in the mobile communication satellite, it is necessary that the frequency error estimating apparatus follows this variation and estimates the frequency error. In other words, in order to increase this level of following the time variation in the Doppler frequency, it is necessary to set the coefficient xcex1 of the multiplier in the averaging filter to a value as large as possible.
As the coefficient xcex1 of the multiplier has been fixed in the conventional frequency error estimating apparatus, there has been a problem that it is difficult to satisfy both increasing the precision in estimating the frequency error and increasing the level of following the time variation in the Doppler frequency of the variation in the Doppler frequency.
It is an object of the present invention to provide a frequency error estimating apparatus and a frequency error estimating method capable of securing high precision in estimating the frequency error in a receiver, without losing the level of following the time variation in the Doppler frequency.
In order to achieve the above object, according to a first aspect of the present invention, there is provided a frequency error estimating apparatus for estimating a frequency error between a local oscillation frequency and a carrier frequency of a received signal in a receiver, the frequency error estimating apparatus comprising: frequency error estimating unit (corresponding to a frequency error estimating unit 11 in an embodiment to be described later) that suppresses a noise component included in the received signal according to a filter coefficient input to a filter, and estimates a frequency error based on an output of the filter; and filter coefficient determining unit (corresponding to a filter coefficient determining unit 12) that calculates a filter coefficient based on a differential of estimate values of the frequency error from a first symbol, and changes the characteristics of the filter.
According to the above aspect, it is possible to satisfy both the level of following the time variation in the Doppler frequency and the precision in estimating the frequency error, by changing the characteristics of a filter for suppressing a noise component of a received signal according to the size of the time variation in the Doppler frequency.
Further, according to a second aspect of the invention, there is provided a frequency error estimating apparatus of the above aspect, wherein the frequency error estimating unit comprises: a modulation component removing unit (corresponding to an M-multiplier 21) that removes a modulation component from the received signal; a phase change information generating unit (corresponding to a D-symbol differential detector 22) that generates phase change information by performing differential detection over a period of a second symbol based on a signal after removing a variation component, an averaging filter unit (corresponding to an averaging filter 23) that suppresses a noise component in the phase change information based on the filter coefficient; and a frequency error estimating unit (corresponding to a coordinate converter 24 and a divider 25) that calculates a frequency error estimate value based on a signal after suppressing the noise component.
According to the above aspect, when the time variation in the Doppler frequency is slow, the filter coefficient is set to a value as small as possible, and when the time variation in the Doppler frequency is fast, the filter coefficient is set to a value as large as possible. The frequency error estimating unit estimates a frequency error based on this filter coefficient. With this arrangement, when the time variation in the Doppler frequency is slow, the filter averaging effect is increased, and it is possible to estimate the frequency error in high precision. On the other hand, when the time variation in the Doppler frequency is fast, it is possible to estimate the frequency error in high precision by following this variation.
Further, according to a third aspect of the invention, there is provided a frequency error estimating apparatus of the above aspect, wherein the frequency error estimating unit comprises: a plurality of differential detection type frequency error estimating units (corresponding to D-symbol differential detection type frequency error estimating units 51a, 51b, . . . , and 51c) that generate phase change information by performing differential detection over periods of predetermined symbols based on a received signal after removing a modulation component, then suppress a noise component in the phase change information based on the filter coefficient, and thereafter estimate a frequency error based on a signal after suppressing the noise component; and a selecting unit (corresponding to a selector 52) that selects an optimum frequency error, based on a predetermined standard, from a plurality of frequency errors that have been estimated by the plurality of differential detection type frequency error estimating unit after performing differential detection over different periods of symbols based on received signals.
According to the above aspect, the frequency error estimating apparatus operates while eliminating the uncertainty in the frequency by selecting an optimum frequency error. Therefore, the differential detection type frequency error estimating unit with a shortest distance of a differential detection symbol determines an estimating range of a frequency error. Then, the differential detection type frequency error estimating unit with a longest distance of a differential detection symbol determines the estimate precision. With this arrangement, it is possible to achieve both the wide estimating range and high estimate precision at the same time.
Further, according to a fourth aspect of the invention, there is provided a frequency error estimating apparatus of the above aspect, wherein the filter coefficient determining unit includes in advance a correspondence table that relates differential values between estimate values of the frequency error to the filter coefficients, whereby, after calculating a differential value, the filter coefficient determining unit, selects a filter coefficient corresponding to the differential value from the correspondence table, and changes the characteristics of the filter according to the value of the selected filter coefficient.
According to the above aspect, there is prepared in advance a correspondence table that relates differential values between estimate values of the frequency error to the filter coefficients. The values of the multiplier coefficients in the correspondence table are determined in advance to take optimum values by a simulation using a calculator or something like that. With this arrangement, it is possible to select easily a filter coefficient corresponding to a differential value by referring to the correspondence table. As a result, it is possible to select an optimum filter coefficient according to a time variation in the Doppler frequency.
Further, according to a fifth aspect of the invention, there is provided a frequency error estimating apparatus of the above aspect, wherein the filter coefficient determining unit further includes a counter of a specific period, whereby the filter coefficient determining unit updates the filter coefficient in a period determined by the counter.
According to the above aspect, while it is possible to update the filter coefficient for each symbol, the updating is carried out in a specific period by using a counter in this case. With this arrangement, it is possible to improve easily the stability of the filter coefficient determining unit.
Further, according to a sixth aspect of the invention, there is provided a frequency error estimating method for estimating a frequency error between a local oscillation frequency and a carrier frequency of a received signal in a receiver, the frequency error estimating method comprising: a first step of suppressing a noise component included in the received signal according to a filter coefficient input to a filter, and estimating a frequency error based on an output of the filter; and a second step of calculating a filter coefficient based on a differential of estimate values of the frequency error from a first symbol, and changing the characteristics of the filter.
According to the above aspect, it is possible to satisfy both the level of following the time variation in the Doppler frequency and the precision in estimating the frequency error, by changing the characteristics of a filter for suppressing a noise component of a received signal according to the size of the time variation in the Doppler frequency.
Further, according to a seventh aspect of the invention, there is provided a frequency error estimating method of the above aspect, wherein the first step comprises: a third step of removing a modulation component from the received signal; a fourth step of generating phase change information by performing differential detection over a period of a second symbol based on a signal after removing a variation component, a fifth step of suppressing a noise component in the phase change information based on the filter coefficient; and a sixth step of calculating a frequency error estimate value based on a signal after suppressing the noise component.
According to the above aspect, when the time variation in the Doppler frequency is slow, the filter coefficient is set to a value as small as possible, and when the time variation in the Doppler frequency is fast, the filter coefficient is set to a value as large as possible. At the first step, a frequency error is estimated based on this filter coefficient. With this arrangement, when the time variation in the Doppler frequency is slow, the filter averaging effect is increased, and it is possible to estimate the frequency error in high precision. On the other hand, when the time variation in the Doppler frequency is fast, it is possible to estimate the frequency error in high precision by following this variation.
Further, according to an eighth aspect of the invention, there is provided a frequency error estimating method of the above aspect, wherein the first step comprises: a plurality of seventh steps of generating phase change information by performing differential detection over periods of predetermined symbols based on a received signal after removing a modulation component, then suppressing a noise component in the phase change information based on the filter coefficient, and thereafter estimating a frequency error based on a signal after suppressing the noise component; and an eighth step of selecting an optimum frequency error, based on a predetermined standard, from a plurality of frequency errors that have been estimated by the plurality of seventh steps after performing differential detection over different periods of symbols based on received signals.
According to the above aspect, the operation is carried out while eliminating the uncertainty in the frequency by selecting an optimum frequency error. Therefore, an estimating range of a frequency error is determined at the seventh step when a distance of a differential detection symbol is shortest. Then, the estimate precision is determined at the seventh step when a distance of a differential detection symbol is longest. With this arrangement, it is possible to achieve both the wide estimating range and high estimate precision at the same time.
Further, according to a ninth aspect of the invention, there is provided a frequency error estimating method of the above aspect, wherein the second step comprises a ninth step of preparing in advance a correspondence table that relates differential values between estimate values of the frequency error to the filter coefficients, whereby, after calculating a differential value, a filter coefficient corresponding to the differential value is selected from the correspondence table, and the characteristics of the filter are changed according to the value of the selected filter coefficient.
According to the above aspect, there is prepared in advance a correspondence table that relates differential values between estimate values of the frequency error to the filter coefficients. The values of the multiplier coefficients in the correspondence table are determined in advance to take optimum values by a simulation using a calculator or something like that. With this arrangement, it is possible to select easily a filter coefficient corresponding to a differential value by referring to the correspondence table. As a result, it is possible to select an optimum filter coefficient according to a time variation in the Doppler frequency.
Further, according to a tenth aspect of the invention, there is provided a frequency error estimating method of the above aspect, wherein the second step further comprises a tenth step for counting in a specific period, whereby the filter coefficient is updated in a period determined by the counter.
According to the above aspect, while it is possible to update the filter coefficient for each symbol, the updating is carried out in a specific period by using a counter in this case. With this arrangement, it is possible to improve easily the stability of the filter coefficient obtained at the second step.